The present invention relates generally to electric power steering apparatus, and more particularly to an improvement in electric power steering apparatus for motor vehicles which impart power of an electric motor to a vehicle""s steering system to reduce a necessary manual steering effort of a human vehicle operator or driver.
Various electric power steering apparatus for motor vehicles are known, in which an electric motor is driven under control of a motor controller unit, on the basis of signals output by a steering torque detector section detecting a steering torque applied to a steering wheel and a vehicle velocity detector section detecting a velocity of the vehicle, so as to reduce a necessary manual steering effort of a human operator. Among the known electric power steering apparatus is one employing a brushless motor for generating a steering torque assist.
The electric power steering apparatus employing a brushless motor can afford a stable steering assist force, because the absence of voltage drops between a brush and a commutator can prevent reduction and variation in motor output power. Further, since the brushless motor presents a smaller moment of inertia than the brush motor, the electric power steering apparatus with such a brushless motor can afford a good steering feel during high-speed straight travel or turning of a steering wheel.
However, with the electric power steering apparatus employing the brushless motor, it is necessary to control an electric current to be supplied to the motor in accordance with a current rotational angle of the motor. Thus, it has been conventional for the electric power steering apparatus to include a rotational detector section for detecting a current rotational angle of the motor and a motor-current detector section, so that the brushless motor is driven in accordance with PWM (Pulse Width Modulation) control on the basis of detection signals output from the rotational detector section and motor-current detector section.
FIG. 5 is a block diagram showing the rotor controller unit for controlling the rotation of the brushless motor. To the brushless motor 101 is connected a VR (variable Reluctance)-type resolver 102 for detecting a current rotational angle of the brushless motor 101.
The motor controller unit 100 for controlling the rotational angle of the brushless motor 101 includes a phase correction section 103, inertia correction section 104 and damper correction section 105.
The phase correction section 103 of the motor controller unit 100 corrects the phase of a steering torque signal T supplied from a steering torque detector section 106 on the basis of a vehicle velocity signal v from a vehicle velocity detector section 107, so as to output an corrected steering torque signal Txe2x80x2 to a target current setting section 108. The inertia correction section 104 generates an inertia correcting signal di, on the basis of the steering torque signal T from the steering torque detector section 106, vehicle velocity signal v from the vehicle velocity detector section 107 and angular-velocity-corresponding signal generated by a differentiation processing section 12Id differentiating a signal corresponding to a rotational angular velocity xcfx89 of a rotor of the motor, outputs the thus-generated inertia correcting signal di to an adder section 109. The damper correction section 105 generates a damper correcting signal dd on the basis of the steering torque signal T from the steering torque detector section 106, vehicle velocity signal v from the vehicle velocity detector section 107 and signal corresponding to the rotational angular velocity xcfx89 of the rotor. The damper correction section 105 outputs the thus-generated damper correcting signal dd to a subtracter section 110.
The target current setting section 108 calculates two-phase target currents Id1 and Iq1 on the basis of the corrected steering torque signal Txe2x80x2 output from the phase correction section 103 and vehicle velocity signal V. The target currents Id1 and Iq1 correspond to a xe2x80x9cdxe2x80x9d axis and xe2x80x9cqxe2x80x9d axis intersecting with the xe2x80x9cdxe2x80x9d axis on a rotational coordinate system synchronized with a rotational magnetic flux produced by a permanent magnet on the rotor of the brushless motor 101. Hereinafter, these target currents Id1 and Iq1 will be referred to as a xe2x80x9cd-axis target currentxe2x80x9d and xe2x80x9cq-axis target currentxe2x80x9d, respectively.
The adder section 109 adds the d-axis target current and q-axis target current Id1 and Iq1 with the inertia correcting signal di, to thereby output inertia-corrected target currents Id2 and Iq2. The subtracter section 110 subtracts the damper correcting signal dd from the inertia-corrected target currents Id2 and Iq2, to thereby output damper-corrected target currents Id3 and Iq3. Hereinafter, these damper-corrected target currents Id3 and Iq3 will be referred to as a xe2x80x9cfinal d-axis target currentxe2x80x9d Id* and xe2x80x9cfinal q-axis target currentxe2x80x9d Iq*, respectively.
The final d-axis target current Id* and final q-axis target current Iq* are passed to an offset calculation section 111, which subtracts d-axis and q-axis detected currents Id and Iq from the final d-axis and q-axis target currents Id* and Iq*, respectively, to thereby calculate offsets DId and DIq and then outputs the thus-calculated offsets DId and DIq to a PI (Proportional and Integral) setting section 112.
The PI setting section 112 performs arithmetic operations using the offsets DId and DIq, to thereby calculate d-axis and q-axis target voltages Vd and Vq such that the d-axis and q-axis detected currents Id and Iq follow the final d-axis target current Id* and final q-axis target current Iq*, respectively. The d-axis and q-axis target voltages Vd and Vq are corrected, via an interference-preventing control section 113 and arithmetic section 114, to d-axis and q-axis corrected target voltages Vdxe2x80x2 and Vdxe2x80x2 that are then delivered to a dq-to-three-phase conversion section 115.
Only one set of the adder section 109, subtracter section 110, offset calculation section 111, PI setting section 112 and arithmetic section 114 are shown in FIG. 5 for purposes of clarity; in practice, however, two separate sets of these sections 109, 110, 111, 112 and 114 are provided for the two target currents Id1 and Iq1.
The interference-preventing control section 113 calculates interference-preventing control correction values for the d-axis and q-axis target voltages Vd and Vq, on the basis of the d-axis and q-axis detected currents Id and Iq and rotational angular velocity xcfx89 of the rotor.
The arithmetic section 114 subtracts the respective interference-preventing control correction values from the d-axis and q-axis target voltages Vd and Vq, to thereby calculate d-axis and q-axis corrected target voltages Vdxe2x80x2 and Vqxe2x80x2 that are output to the dq-to-three-phase conversion section 115.
The dq-to-three-phase conversion section 115 converts the d-axis and q-axis corrected target voltages Vdxe2x80x2 and Vqxe2x80x2 to three-phase target voltages Vu*, Vv* and Vw* and outputs the thus-converted three-phase target voltages Vu*, Vv* and Vw* to a motor drive section 116.
The motor drive section 116 includes a PWM-controlled voltage generation section and inverter circuit (both not shown). The motor drive section 116 generates, by means of the not-shown PWM-controlled voltage generation section, PWM-controlled voltage signals UU, VU and WU corresponding to the three-phase target voltages Vu*, Vv* and Vw*, and it outputs these PWM-controlled voltage signals UU, VU and WU to the not-shown inverter circuit. Then, the inverter circuit generates three-phase A.C. driving currents Iu, Iv and Iw corresponding to the PWM-controlled voltage signals UU, VU and WU, which are supplied via three-phase driving current paths 117 to the brushless motor 101. The three-phase A.C. driving currents Iu, Iv and Iw are each a sine-wave current for driving the brushless motor 101 on the basis of the PWM control (i.e., PWM driving of the brushless motor 101).
Motor current detector sections 118 and 119, which are provided on two of the three-phase driving current paths 117, detect two driving currents Iu and Iw among the three-phase A.C. driving currents Iu, Iv and Iw to be supplied to the brushless motor 101 and outputs the detected driving currents Iu and Iw to a three-phase-to-dq conversion section 120. In turn, the conversion section 120 calculates the remaining driving current Iv on the basis of the detected driving currents Iu and Iw, and it converts these three-phase A.C. driving currents Iu, Iv and Iw into d-axis and q-axis detected currents Id and Iq.
Signals sequentially generated by the resolvers 102 are supplied in succession to an R/D (resolver/digital) conversion section 121. The R/D conversion section 121 calculates an angle (rotational angle) xcex8 of the rotor relative to the stator of the brushless motor 101 and then supplies the dq-to-three-phase conversion section 115a and three-phase-to-dq conversion section 120 with a signal corresponding to the calculated rotational angle xcex8. Further, the R/D conversion section 121 calculates a rotational angular velocity xcfx89 of the rotor relative to the stator of the brushless motor 101 and then supplies the damper correction section 105, differentiation processing section 121d and interference-preventing control section 113 with a signal corresponding to the calculated rotational angular velocity xcfx89. The VR-type resolver 102 and RD conversion section 121 together constitute a rotational detector section 102A for detecting a rotational angle of the brushless motor 101.
As illustrated in FIG. 6, all the components, except for the various sensors and inverter circuit, of the motor controller unit 100 are incorporated in an electronic circuitry unit, and in fact, the motor controller unit 100 is implemented by a microcomputer 122; that is, the respective functions of the components are performed by processing based on software programs.
In FIG. 6, an interface circuit 123 includes an A/D converter that converts, into digital representation, the steering torque signal T from the steering torque detector section 106, vehicle velocity signal v from the vehicle velocity detector section 107 and engine rotation signal r from an engine rotation detector section 124. The digital signals thus converted in the interface circuit 123 are passed to the microcomputer 122.
Another interface circuit 125 converts, into digital representation, the driving currents Iu and Iw detected by the motor current detector sections 118 and 119 and delivers the thus-converted digital signals to the microcomputer 122. Still another interface circuit 126 passes an exciting current from an R/D converter 127 to the resolver 102 and an output signal of the resolver 102 to the R/D converter 127. As will be described later, the R/D converter 127 generates an angle signal on the basis of the output signal of the resolver 102 and sends the thus-generated angle signal to the microcomputer 122. The motor drive section 116 includes a pre-drive circuit 128 and an inverter circuit having six power FETs.
External crystal oscillator 129 and capacitors 130 and 131 are connected to the microcomputer 122, and the microcomputer 122 divides an oscillation frequency of the crystal oscillator 129 to generate a frequency fPWM of PWM signals for driving the brushless motor 101 (hereinafter also called a xe2x80x9cPWM driving frequencyxe2x80x9d).
Further, a crystal oscillator 132 and capacitors 133 and 134 are connected to the R/D converter 127, and the R/D converter 127 divides an oscillation frequency of the crystal oscillator 132 to generate a frequency fRES of exciting signals to be sent to the resolver 102.
Generally, in order to provide an electric power steering apparatus capable of affording a smooth steering feel, smooth outputs of a brushless motor are required. For this purpose, the motor controller unit may perform vector control on the brushless motor on the basis of the output signals of the motor rotation detector section and motor current detector as set forth above and supplies sine-wave currents to the brushless motor as motor currents so that the motor produces outputs with small torque variations.
Specifically, the sine-wave currents are supplied to the brushless motor via the motor drive (inverter) section composed of switching elements, such as FETs, and peripheral circuits associated therewith. Such switching elements are driven at the PWM driving frequency fPWM beyond the audible range and thereby supplies driving power to the brushless motor 101.
Further, because the vector control requires detection of an absolute rotational angle of the brushless motor 101, the electric power steering apparatus includes a rotational detector section, such as a resolver, for detecting a rotational angle, angular velocity, angular acceleration, etc. of the motor 10. The resolver detects gap variations of an iron core of the rotor to thereby detect a rotational angle of the motor.
FIG. 7 is a diagram explanatory of the operating principles of the resolver. Coil A is provided adjacent to one side of the rotor R as an exciting coil, while coils B and C are provided adjacent to the opposite side of the rotor R as two output coils forming a right angle therebetween. Magnetic field produced by a current flowing through the energizing coil A flows in the output coils B and C. With the current varying over time, an inductive electromotive force is produced in the output coils B and C in accordance with the Faraday""s law of electromagnetic induction.
Namely, a voltage of an angular frequency xcfx89E as represented by Mathematical Expression (1) below is applied to terminals R1 and R2 of the energizing coil A, as single-phase excitation.
ER1-R2=E sin xcfx89Etxe2x80x83xe2x80x83Mathematical Expression (1) 
Thus, when the rotor R is at an angle xcex8, a voltage as represented by Mathematical Expression (2) below is output from terminals S1 and S3 of the output coil B while a voltage as represented by Mathematical Expression (3) is output from terminals S2 and S4 of the output coil C.
Es1-s3=KE sin xcfx89Etxc3x97cos xcex8xe2x80x83xe2x80x83Mathematical Expression (2) 
Es2-s4=KE sin xcfx89Etxc3x97sin xcex8xe2x80x83xe2x80x83Mathematical Expression (3) 
FIG. 8 is a block diagram explanatory of the RD conversion principles of the R/D converter 127. The voltage Es1-s3 input to the R/D converter 127 is supplied to an arithmetic section 135, which calculates a product between the input voltage Es1-s3 and a sine value of an angle xcfx86 (sin xcfx86) stored in an internal ROM. Similarly, the voltage Es2-s4 input to the R/D converter 127 is supplied to another arithmetic section 136, which calculates a product between the input voltage Es2-s4 and a cosine value of the angle xcfx86 stored in the internal ROM. Then, an arithmetic section 137 determines a difference D1 as represented by Mathematical Expression (4) below.                                                         D1              =                                                                    E                                          s2                      -                      s4                                                        xc3x97                  cos                  ⁢                                      xe2x80x83                                    ⁢                  φ                                -                                                      E                                          s1                      -                      s3                                                        xc3x97                  sin                  ⁢                                      xe2x80x83                                    ⁢                  φ                                                                                                        =                              KE                ⁢                                  xe2x80x83                                ⁢                sin                ⁢                                  xe2x80x83                                ⁢                                  ω                  E                                ⁢                t                xc3x97                                  (                                                            sin                      ⁢                                              xe2x80x83                                            ⁢                      θ                      ⁢                                              xe2x80x83                                            ⁢                      cos                      ⁢                                              xe2x80x83                                            ⁢                      φ                                        -                                          cos                      ⁢                                              xe2x80x83                                            ⁢                      θ                      ⁢                                              xe2x80x83                                            ⁢                      sin                      ⁢                                              xe2x80x83                                            ⁢                      φ                                                        )                                                                                        Mathematical        ⁢                  xe2x80x83                ⁢        Expression        ⁢                  xe2x80x83                ⁢                  (          4          )                    
The thus-determined difference D1 is modified as represented Mathematical Expression (5) below.
D1=KE sin xcfx89Etxc3x97sin(xcex8xe2x88x92xcfx86)xe2x80x83xe2x80x83Mathematical Expression (5) 
Synchronized detector section 138 detects a signal indicative of the difference d1 in synchronism with an exciting input voltage, so that a signal D2 as represented Mathematical Expression (6) below is output from the synchronized detector section 138.
D2=sin(xcex8xe2x88x92xcfx86)xe2x80x83xe2x80x83Mathematical Expression (6) 
The signal D2 (sin(xcex8xe2x88x92xcfx86)) is passed to a VCO (Voltage-Controlled Oscillator) section 139 and counter 140, which output an anglexcex8 by increasing or decreasing the value of the angle xcex8 such that the signal D2 always takes a zero value.
In short, a single-phase input voltage is excited by the sine wave represented by Mathematical Expression (1), and two-phase (sine and cosine) output voltages, modulated with the sine and cosine waves represented by Mathematical Expression (2) and Mathematical Expression (3), are obtained. Then, the two-phase outputs are subjected to the above-mentioned R/D conversion to thereby provide an angle output. Here, the exciting frequency fRES is approximately 10 kHz.
In this case, if switching noise caused by the PWM driving is introduced in the outputs of the resolver, the outputs of the R/D converter 127 will present variations corresponding to a difference between the PWM driving frequency fPWM and the exciting frequency fRES (f1=fPWMxe2x88x92fRES) or difference between respective harmonics (higher-order frequency components) of the PWM driving frequency fPWM and the exciting frequency fRES (f2=nxc3x97fPWMxe2x88x92mxc3x97fRES where n=1, 2, . . . , m=1, 2, . . . ). As a consequence, the outputs of the brushless motor too will present variations corresponding to the frequency difference f1 (Hz) or f2 (Hz). In the conventional motor controller unit of FIG. 6, such output variations would result for the following reason even if the PWM driving frequency fPWM and exciting frequency fRES are set to be identical to each other.
Namely, in the conventional motor controller unit of FIG. 6, the frequency fPWM of the PWM signal (i.e., PWM driving frequency fPWM) is generated by the microcomputer 122 dividing the oscillation frequency of the crystal oscillator 129, and the frequency fRES of the exciting voltage (i.e., exciting frequency fRES) is generated by the R/D converter 127 dividing the oscillation frequency of the crystal oscillator 132. Because the PWM driving frequency fPWM and exciting frequency fRES are generated on the basis of two separate crystal oscillators, there would be caused variations in load capacitance due to individual differences etc. between the crystal oscillators 129 and 132 and between the capacitors 130, 131 and 133, 134. Thus, even if two crystal oscillators of stable frequencies are employed, there would be caused a frequency difference between signals generated from the two different crystal oscillator circuits. For this reason, it has been difficult to make the PWM driving frequency fPWM and exciting frequency fRES exactly identical to each other. Therefore, the R/D converter 127 produces considerable variations in its outputs, which would lead to undesired variations or fluctuations in the steering assist force imparted by the brushless motor 101. The variations in the steering assist force would cause vibrations of the steering wheel, thereby significantly impairing the steering feel.
In view of the foregoing prior art problems, it is an object of the present invention to provide an electric power steering apparatus of a type employing a brushless motor where there occurs no impairment of a steering feel due to wear and tear of a motor brush and moment-of-inertia of a motor rotor and which can afford a smooth steering feel with minimized variations in the steering assist force by reducing variations in the output power of the brushless motor.
In order to accomplish the above-mentioned object, the present invention provides a n electric power steering apparatus which comprises: a steering torque detector section for detecting steering torque applied to a steering wheel; a brushless motor for imparting a steering torque assist to a steering system; a rotational angle detector section for detecting a rotational angle of the brushless motor, the rotational angle detector section including a resolver; a motor current detector section for detecting a current supplied to the brushless motor; and a motor controller unit for controlling PWM driving of the brushless motor at a predetermined PWM driving frequency, on the basis of output signals of at least the steering torque detector section, rotational angle detector section and motor current detector section. In this electric power steering apparatus, one of the predetermined PWM driving frequency and the predetermined exciting frequency of the resolver is set to be an integral multiple of the other of the predetermined PWM driving frequency and the predetermined exciting frequency.
Because one of the predetermined PWM driving frequency at which the brushless motor is driven and the predetermined exciting frequency of the resolver is set to be an integral multiple of the other, the difference between the PWM driving frequency fPWM and the exciting frequency fRES (i.e., f1=fPWMxe2x88x92fRES) or the difference between respective harmonics (higher-order frequency components) of the PWM driving frequency fPWM and the exciting frequency fRES (i.e., f2=nxc3x97fPWMxe2x88x92mxc3x97fRES where n=1, 2, . . . , m=1, 2, . . . ) can be 0 Hz, or the difference f1 or f2 can be an integral multiple of the exciting frequency fRES. Thus, the outputs from the R/D conversion section and brushless motor present no substantial variation, so that variation-free steering assist force can be applied to the steering wheel. As a result, the present invention can afford a smooth steering feel.
In a preferred implementation, the electric power steering apparatus employs a same oscillator for generating predetermined signals, and both the PWM driving of the brushless motor and excitation of the resolver are performed in accordance with the output signals of the same oscillator in such a manner that one of the predetermined PWM driving frequency and the predetermined exciting frequency of the resolver is set to be an integral multiple of the other of the predetermined PWM driving frequency and the predetermined exciting frequency. Because the predetermined PWM driving frequency fPWM at which the brushless motor is driven and the predetermined exciting frequency exciting frequency fRES of the resolver are generated on the basis of the output signals of the same oscillator and one of the predetermined PWM driving frequency and the predetermined exciting frequency of the resolver is set to be an integral multiple of the other, the difference between the PWM driving frequency fPWM and the exciting frequency fRES (i.e., f1=fPWMxe2x88x92fRES) or the difference between respective harmonics (higher-order frequency components) of the PWM driving frequency fPWM and the exciting frequency fRES (i.e., f2=nxc3x97fPWMxe2x88x92mxc3x97fRES where n=1, 2, . . . , m=1, 2, . . . ) can be reliably set to be 0 Hz, or the difference f1 or f2 can be reliably set to be an integral multiple of the exciting frequency fRES. Thus, the outputs from the R/D conversion section and brushless motor present no substantial variation, so that variation-free steering assist force can be applied to the steering wheel. As a result, the present invention can afford a smooth steering feel.